The temperature affects the function of electronic circuits e.g. owing to temperature dependence of the voltage of semiconductor junctions. In amplifiers a temperature change causes changes in currents and as a result of that change in the gain. Concerning power stage of a radio transmitter a change in the gain means an unintentional change of the antenna's radiation power. In mobile terminals the transmitting power is intentionally tried to set to a value, which is just sufficient. Unnecessary high transmitting powers cause rise of the noise level in radio network, which is tried to be avoided, of course. Especially in WCDMA system (Wideband Code Division Multiple Access), which is becoming more general, it is important to keep the noise level in the network as low as possible. Hence the gain of the transmitters of terminals accurately has to stick to the set value.
Drift caused by temperature changes has for a long time been prevented by compensation principle. In that case a circuit is designed so that the effects of temperature sensitive circuit elements to the quantity at issue are balanced. For example the voltage over a series connection of a forward biased semiconductor junction and a 6.8 volt zener diode is nearly independent of the temperature, because the voltage of a forward biased semiconductor junction changes about −2 mV/° C. and the voltage of a zener diode about +2 mW/° C. In amplifiers a change in current and in level of the output signal, caused by temperature changes, can be prevented by arranging an opposite change, which is based on change in a certain junction voltage. FIG. 1 shows an example of known variable gain amplifier with temperature compensation. The amplifier 100 comprises an amplifier proper 110 and control circuit 120 thereof. The amplifier proper 110 has a differential pair Q1-Q2. The emitters of these transistors are connected to a controllable signal current source 111, the second terminal of which being connected to the ground. The collector of the first transistor Q1 is directly connected to the supply voltage VS and the collector of the second transistor Q2 is connected to a load of the amplifier, the load being not shown in FIG. 1. When the base currents of the transistors are not taken into account, the current iin of the current source 111 is sum of the first collector current i1 and the second collector current i2. The signal current source 111 is controlled by the input signal vin of the amplifier. The second collector current i2 is at the same time the output signal iout of the amplifier. The base of the first transistor Q1 is connected to a reference voltage Vref through a resistor R11 and the base of the second transistor Q2 is connected to the output of the control circuit 120 through a resistor R12.
The term “differential pair” means in this description and in the claims two transistors, the emitters of which are connected together. The total emitter current is then divided between the transistors in a certain ratio depending on the control led to the bases.
The control circuit 120 comprises an operational amplifier A11, a feed back resistor R13 thereof and the second current source 121. The non-inverting input of the operational amplifier is connected to the above-mentioned reference voltage Vref and the current source 121 is connected from the inverting input to the ground. The current source is controlled externally by the gain control signal G. The direction of the source current IGT is towards the ground, in which case the output voltage of the control circuit VGT=Vref+R13·IGT. So the minimum value of the voltage VGT to be led to the base of the second transistor Q2 is Vref, or same as the base voltage of the first transistor Q1. The currents i1 and i2 of the differential pair Q1-Q2 are in that case equal in amount, whereupon the current gain GI=iout/iin=0,5. When the source current IGT is enhanced by the control signal G, the voltage VGT is becoming higher according to the expression above. This results in that the current i2 of the second transistor is enhanced and the current i1 of the first transistor is reduced the same amount. The current iin remains unchanged, whereupon the current gain GI becomes greater than 0,5. The maximum value of the current gain is one, in which case the current iin of the current source 121 flows wholly through the second transistor Q2.
So the gain is controlled by the control signal G. However the value of the gain value is in principle affected also by the temperature. A temperature change causes a change in the difference between control voltages of the differential pair Q1-Q2, or in offset-voltage, and a corresponding change in collector currents i1 and i2. The direction of the change is such that the current i2 and the current gain are reduced when the temperature rises. In the same way a temperature change causes a change in the offset-voltage of the operational amplifier A11, which results in that the values of the collector current i2 and the current gain are changed. To compensate the effect of the temperature, the current source 121 has a built-in temperature dependence: When the temperature rises, the source current IGT is enhanced proportionally to the absolute temperature so that the effect on the gain is equal in amount but opposite, compared with the effect of the changes of the offset-voltages.
A flaw of the above-described arrangement is, that the control range of the gain with it's temperature compensation is only half of the theoretic maximum range. In theory the output current iout can be varied within the range 0 . . . iin, corresponding control range 0 . . . 1 of the gain. This could be obtained by arranging the reference voltage, used by the control circuit 120, to be e.g. 0.2 V lower than the reference voltage used by the amplifier proper 110. However, in that case the temperature compensation would not function when the current gain is less than 0.5. On the contrary, a change in the source current IGT would only enhance an error produced inside the differential pair.